1. Reference should be made to the circuit diagram at the end of this


2. A 75-ohms unbalanced aerial source is connected to the tuned r.f.
amplifier trough a three-section 30 Mc/s low-pass filter and a five-
position attenuator covering a range of 0 to 40 dB. Switch S2 selects wide-
band 75-ohms or wideband (high impedance) or any one the five double-tuned
aerial coils L4-L8 for tuned operation. These aerial coils are aligned by
means of dust iron cores. The aerial is tuned by a capacitor C18A/B which
is switched out of circuit in both wideband positions.


3. The incoming signal is fed via C28 and grid stopper R25 to the grid
of V3B; the r.f. stage (V3) employs a variable-mu, low-noise double-
triode; the two halves of the valve are connected in cascode so as to utilize
the low-noise high-gain properties of the valve. A delayed a.v.c. voltage,
derived from a shunt diode network, is applied to the grid of V3B when the
signal level is approximately 10µV. The capacitors C40 and C41 ensure that
the cathode is adequately decoupled over the wide frequency range. Ferrite
beads have been fitted to the heater lead, connected to pin 4, the anode of
V3A and the cathode of V3B adjacent to C41, to prevent parasitic oscillations


4. The amplified signal is passed to a 30 Mc/s low-pass filter which has
a substantially flat responseover the frequency range. L27, C47 and
R28 constitute the first 'L half Section' of the filter. The signal is then
fed at low impedance (680-ohms) trough the coupling capacitor C74 and the grid
stopper R45 to the control grid of V7, the first mixer stage. The input
capacitance of V7 forms the capacitance to chassis betweeenL15 and L17 required
to the filter network.

NOTE: This capacitance is not critical, therefore no adjustment will be
necessary should V7 be changed.


5. This circuit comprises a cathode-coupled Hartley oscillator stage (V5)
which may be continously tuned over the frequency range of 40.5 to
69.5 Mc/s. The frequency determining components are an inductance L36 and a
variable capacitanceC76. Alignment is accomplished by adjusting aluminium core
of L36 and the trimming capacitor C77. The variable capacitor C76 is coupled
to the Mc/s dial which is calibrated from 0 to 29 Mc/s. The anode load consists
of L20, a compensating inductance which is wound on a 470-ohm resistor R18.
The oscillator is coupled via C85 to the signal grid of the first mixer stage
V7 and also via C42 to the control grid of the harmonic mixer V4.

NOTE: The Mc/s dial calibration may be affected if V5 if changed. The
necessary correction may be made by adjusting C77 with the Mc/s dial
set to 29 Mc/s.


6. the outputs from the 30 Mc/s low-pass filter and the variable frequency
oscillator VFO-1 are fed to the signal grid of the mixer stage (V7)
which produces a signal at 40 Mc/s. The signal is then passed to a 40 Mc/s
band-pass filter which forms the anode load of this stage.


7. The 40 Mc/s band-pass filter consists of eight over-coupled tuned
circuits connected in cascade and is tuned by the trimming capacitors
C21, C33, C43, C53, C61, C70, C79 and C88. This filter, which has a passband
of 40 Mc/s ±650 kc/s, ensures that only the required 1 Mc/s spectrum of
signals is passed to the second stage. This filter is deliberately set to a
slightly wider passband than is theoretically required, to allow for possible
drift in VFO-1.


8. The frequency of the crystal oscillator V1 may be set precisely to
1 Mc/s by adjusting the trimming capacitor C2A. The crystal XL1 which
is connected between the control grid and the screen grid is electron coupled
to the anode. The anode coil L2 is adjusted to resonate at 1 Mc/s by means of
a dust iron core. The fixed capacitors C9, C10 and C11 complete the tuned
circuit. When an external signal is applied to socket SK3, the valve operates
as an amplifier.

9. The output from V1 is capacitance-coupled to the harmonic generator
V2 and via SK2 to a "T" adptor for feeding a 1 Mc/s input into the
l.f. converter and also the control grid of the mixer valve V13.


10. The 1 Mc/s signal is fed via coupling capacitor C8 to the control grid
of the harmonic generator V2. The h.t. is fed to the screen grid via
R12 and is decoupled by C8A. Harmonics produced at this stage are passed to a
32 Mc/s low-pass filter.


11. The megacycle harmonics are fed trough a 32 Mc/s low-pass filter
circuit to prevent harmonics other than those required from passing
to the harmonic mixer (V4). Limited control over the cut-off frequency is
provided by C7 which is adjusted to equalize the output from yhe filter at
the frequencies corresponding to 28 and 29 Mc/s on the MEGACYCLE dial.


12. The outputs from the 32 Mc/s low-pass filter and VFO-1 are mixed in
the harmonic mixer by applying the filtered megacycle harmonics to
the suppressor grid and the output from the VFO-1 to the control grid. The
37.5 Mc/s output is selected by the tuned anode load, consisting of a fixed
capacitor C50 and an inductance L28 which may be adjusted by means of a dust
iron core, and coupled by C51 to V6. R36 is grid stopper.


13. The anode load of V6 is a tuned circuit consisting of a fixed capacitor
C67 and an inductor L33 Which is tuned to 37.5 Mc/s. Frequency
adjustment is by the dust iron core L33. This stage feeds the amplified signal
via C68 to the following stage V8. The 37.5 Mc/s signal is then passed to the
37.5 Mc/s band-pass filter. The anode load of this stage is provided by this


14. The 37.5 Mc/s band-pass filter consists of eight under-coupled tuned
circuits arranged in cascade. These filter sections may be tuned by
C24, C35, C45, C55, C63, C72, C81 and C91 respectively. This filter, which
has a passband of 300 kc/s, allows for possible drift in VFO-1. The narrow
passband and high rejection to frequencies outside the passband prevent
spurious signals from reaching the second mixer stage (V9).


15. The filtered 37.5 Mc/s signal is further amplified by V10 before being
passed to the second mixer stage (V9). To prevent interaction between
the 40 Mc/s band-pass filter and the 37.5 Mc/s tuned circuit (L50 and C113)
and to enable either circuit to be adjusted without affecting the other, a
balancing circuit is included which is shown in simplified form in fig.4. The
40 Mc/s signal is introduced into the 37.5 Mc/s tuned circuit at a point of
zero r.f. potential since L50 is centre tapped and C108 is adjusted to be
equal to the total of the capacitance of V10 anode to chassis. C107 and the
input capacitor of V9.

NOTE: The anode load of V10 is adjusted to 37.5 Mc/s by adjusting the dust
iron core in L50. The balancing circuit will be affected if V9 or V10
is changed.


16. This mixer (V9) produces the second intermediate frequency of 2-3 Mc/s
by mixing the 40 Mc/s i.f. and the 37.5 Mc/s signal. The tuned circuit
formed by L300, C300 remove the 37.5 Mc/s frequency whilst the other tuned
circuit formed by L301, C301 remove the 6 Mc/s frequency so that only the
second i.f. is passed to the 2-3 Mc/s band-pass filter preceding the third


17. This filter consists of two pre-tuned band-pass filter sections. The
characteristic impedance of the filteris 1000-ohms.


18. The output from the 2-3 Mc/s band-pass filter is resistance-capacitance
coupled to the signal grid of V25 together with the output (3.6-4.6
Mc/s) from the second v.f.o. amplifier V11 when the V.F.O. switch (S300) is set
to the INT. position. With the V.F.O. switch set to the EXT. position, V11
operates as a buffer amplifier. This mixer (V25) produces the third intermediate
frequency of 1.6 Mc/s. The signal is then fed to a 1.6 Mc/s band-pass filter
which forms the anode load of this stage.

19. The 1.6 Mc/s band-pass filter consists of two double-tuned i.f. trans-
formers, the first section of the filter is formed by C320, L306, L309
and C325 and the second section by C332, L313, L314, C334. This filter has a
bandwidth of 13 kc/s.


20. The second variable frequency oscillator, covering a frequency range
3.6 to 4.6 Mc/s, is an electron coupled Hartley circuit embloying one
half of double-triode V12. The oscillator frequency is determined by an
inductance L55, two fixed capacitors C303, C305, a trimming capacitor C306 and
a variable capcitor C301. The KILOCYCLES scale which is calibrated between 0
and 1000 kc/s is coupled to this variable capacitor.

21. The output from VFO-2 is resistance-capacitance coupled to the grid of
V12A, a cathode-follower stage. With the V.F.O. switch set to the INT.
position the output from V12A is fed via PL305 and PL300A to the control grid
of the second v.f.o. amplifier V11. In the EXT. position the external 3.6 to
4.6 Mc/s signal is fed ti V11.


22. the output from the 1.6 Mc/s band-pass filter is directly coupled to
the signal grid of a pentagrid valve V26; it is mixed with a 1.7 Mc/s
signal from V27 fed via the coupling capacitor C339 to the oscillator grid of
V26. The resistor R68 completes the d.c. path from this grid to earth. The
100 kc/s output from this mixer stage is then fed via SK6, PL6 to the crystal
filter unit.


23. The frequency from the crystal oscillator C27 may be set precisely to
1.7 Mc/s by adjusting the trimming capacitor C337. The crystal XL300
which is connected between the control grid and the screen grid is electron
coupled to the anode. When an external signal is applied to socket SK303A the
valve operates as an amplifier. The output from this circuit is fed via C339
to the oscillator grid of the fourth mixer V26.


24. Six alternative switched i.f. bandwidths are available as follows:-

100 c/s ) Crystal 1.2 kc/s )
300 c/s ) 3.0 kc/s )
6.5 kc/s ) L - C
13.0 kc/s )

25. In the crystal positions the fourth mixer anode is connected to L48
in the crystal filter. L47 and L49 provide a balanced output which is
tuned by capacitors C109 and C110. In the 100 c/s position, the balanced
output is connected via crystals XL2 and XL5 to the first tuned section of the
100 c/s L-C filter. The differential trimmer C118 is the phasing control for
this bandwidth. XL3, XL6 the capacitor C119 form a similar circuit for the
300 c/s position. Damping resistors R64 and R65 are connected across the tuned
circuits to obtain the required bandwidth.


26. This filter consists of four tuned circuits arranged in cascade. In the
L-C bandwidth positions, the signal is fed to the tuned circuit formed
by L61 and the combination of the capacitors C145, C146, C146A and C147. The
second section consists of L62 and L63 in series with C152, C152A and C153. The
final section consisting of L68 and L71 in series with C161 and C162, is damped
by the series resistors R86, R87A and R88 according to the bandwidth. In the
L-C positions the output is taken from a capacitive divider formed by C161 and
C161A with C170, to equalize the gains in the L-C and crystal bandwidth

27. The L-C banwidths are obtained by varying the degree of coupling
between each section of the filter in addition to the damping resistors
in the final stage. The capacitor C175 is included to compensate for the
effective reduction of the input capacitance of V14, appearing across the tuned
circuit, when switching from crystal to L-C positions.

28. To maintain the input capacitance of the L-C filter, in the crystal
positions, a trimming capacitor C148 is switched into circuits. This
trimmer is adjusted to be equal to the output capacitance of V26 and the
screened cable. In the crystal bandwidth positions, the L-C filter is operating
in its narrow bandwidth positions, i.e. 1.2 kc/s.

NOTE: The 470-kilohm damping resistors R77 and R80 are disconnected
except during filter alignment.


29. The output from the L-C filter is passed trough a coupling capacitor
C164 to the control grid of the pentode amplifier valve V14. This grid
is returned via R96 to the a.v.c. line which is filtered at this point by R102
and C173. The screen potential is derived from a potential divider formed by
R93, R97 and RV4. This stage is coupled to the second i.f. amplifier and the
i.f. output stage by a double tuned transformer having an over-coupled


30. The signal from the first i.f. tranformer is fed trough the grid
stopper R114 to the control grid of the second i.f. amplifier. H.T. is
supplied to the screen via the dropping resistor R113 and is decoupled by C181.
The anode load is tuned circuit consisting of L77, C192 and C191. This circuit
is heavily damped by R112. The secondary winding L78 and L79 is tuned by C195
and C195B with R120A as a damping resistor. The output is fed to the diode
detector anode.


31. The low potential end of L79 is connected through the r.f. filter (C209,
R128, C210, C219 and C211) to the diode load R130. With the meter
switched to R.F. LEVEL, the meter indicates the detector diode current. The
resistor R131 is incluced to complete the diode detector circuit when the meter
is switched out of circuit.


32. The noise limiter diode (pins 2 and 5 of V21) is connected in a series
circuit to operate at approximately 30% modulation. its operation is
explained with reference to Fig.5.

33. The d.c. path from point A is trough R134, R135, the diode and R137.
The a.f. signal path from detector diode load is through C216, the
diode and C218 when S8 is open. In the presence of a signal, a negative
potential varying with the depth of modulation, will be developed at point A
thus causing the diode to conduct. The negative potential at B, will be lower
than at A and will be maintained at a constant level due to the long time
constant of R134 and C217. R135 allows the cathode potential to vary in
sympathy with the modulation provided the modulation depth does not exceed 30%.
The potential appearing at the cathode of the noise limiter diode therefore
consists of a steady negative potential with the modulation superimposed. When
noise impulses corresponding to high modulation peaks appear at point A and via
C216 at point C, the voltage across the diode changes sign thereby causing the
diode to stop conducting and open-circuit the a.f. signal path. With S8 in the
OFF position the limiter is inoperative.


34. The signal appearing at the anode of V16 is passed through the capacitor
C139 to the anode of the a.v.c. diode. The diode load is formed by R116.
A positive potential derived from R120, R121 and R122, supplies the required
a.v.c. delay voltage to the cathode of this diode.When A.V.C. switch is in the
SHORT position and the SYSTEM switch set to a position in which the a.v.c. is
operative, i.e. A.V.C., CAL. or CHECK B.F.O., the anode of the a.v.c. diode is
connected to the a.v.c. line via L81 and R127. The choke L81 is tuned by C203
to a frequency slightly below 100 kc/s so that is presents a small capacitance
at 100 kc/s, thus R127 is prevented from shunting the diode load. When the
signal level falls, the capacitors C182 and C173 discharge through R118, R127
and L81 into the diode load resistor R116. The a.v.c. potential is brought out
via R123 to the tag strip at the rear of the receiver for external use if
required. With the SYSTEM switch set to the MANUAL position, the a.v.c. line is
connected to the R.F./I.F. GAIN control RV1, thus the gain of the 100 kc/s
amplifiers may be varied by adjusting the negative potential applied to the
a.v.c. line.


35. Audio frequencies are applied to the control grid of V23B via RV2 the
A.F. GAIN control. The output transformer (T2) provides four separate
outputs as follows: 1W into 3-ohms, and three windings supplying 3mW into 600-

36. The internal loudspeaker (which may be switched out of circuit by
operating S11) is connected across the 3-ohm winding. The headphone
jacks JK1 and JK2 are connected across one of the 600-ohms windings.


37. The audio frequencies are also applied to the grid of V23A via RV3, the
A.F. GAIN LEVEL control; this control presets the level from output
transformer T3. The transformer provides a 10mW output at 600-ohms which is
suitable for direct connection to landlines. A bridge rectifier MR1 is connected
across the output via R142 and R143. Th meter may be switched across the
rectifier circuit so that the operator can monitor the a.f. output.


38. The beat frequency oscillator (V19) employs an electron-coupled Harley
circuit. The oscillation frequency is determined by a fixed inductor
L82 and a variable capacitor C200 in parallel with C202 and C201. the trimming
capacitor C201 is adjusted to produce an output frequency of preisely 100 kc/s
when the beat frequency oscillator frequency control is set to zero. Bias is
applied to this valve by C199 and R125.

39. The b.f.o. output is coupled to the diode detector anode via C215. The
b.f.o. is supplied with h.t. via S7 except when SYSTEM switch is in the
CAL. or STANDBY positions.


40. The control grid of V17 is connected to the secondary of the first
100 kc/s i.f. transformer which feeds the stage with the 100 kc/s
signal. The screen resistor R108 and the cathode bias resistor R115 are of the
same values as used in the scond 100 kc/s i.f. amplifier, hence the a.v.c.
characteristic of this stage is identical to that of the main receiver. The
anode load resistor R109 feeds the auto transformer L76 via blocking capacitor
C189. This transformer provides a 70-ohms output at PL8 and PL9 for external

NOTE: PL8 and PL9 are connected in parallel, therefore only one 100 kc/s
output is available at 75-ohms, and to avoid a mismatch the other
connection should be made at high impedance.


41. The crystal calibrator, controlled by the 1 Mc/s crystal or by the
1 Mc/s standard input to V1, feeds signals at 100 kc/s intervals to
the signal grid of the third mixer stage to provide calibration check points.
The calibration can only be carried out when the V.F.O. switch S300 is set
to the INT. position.

42. The 1 Mc/s signal, fed through SK2, is connected through PL2 and the
grid stopper R83 to the first grid of the mixer valve V13. The anode
load consists of a 100 kc/s tuned circuit (L70, C167) and is coupled to the
control grid of V15 through the capacitor C168. The anode load of V15 (L75,
C117) is tuned to 900 kc/s and is coupledvia C178 to the third grid of V13.
V15 is heavily biased so that it functions as a frequency multiplier.

43. An output of 900 kc/s, appearing across the tuned circuit (L75, C177)
is coupled to grid 3 of V13 thereby producing a difference frequency
of 100 kc/s relative to the 1 Mc/s input. The 100 kc/s output appears across
the anode tuned circuit (L70, C167) and is fed to the control grid of V15. The
ninth harmonic is selected in turn by the anode tuned circuit (L75, C177) of
V15 and fed back to the third grid of V13 to provide the beat frequency of
100 kc/s with the 1 Mc/s input. This crystal controlled regenerative circuit
is thus self-maintaining. The 100 kc/s output is obtained from the coil L69
which is mutually coupled to L70 and fed via the octal plug (PL7) to the
cathode-follower V12A.


44. The primary of the mains transformer is tapped to provide for inputs
of 100-125 and 200-250V. To remove mains-borne interference the
capacitors C224 and C225 are incluced. The secondary winding of T1 feeds a
bridge-connected full-wave rectifier MR4, MR5, MR6 and MR7 whose output is
filtered by C206, L80 and C198 and fed via the receiver muting relay RL1/1
to the SYSTEM switch S5. A 120-ohm resistor R124 is connected between the
negative line and earths thus providing a negative 25V d.c. supply for gain
control purposes.


45. The following conditions exist for each setting of the SYSTEM switch.
The link on h.t. adaptor terminal is assumed to be in position.

(1) STANDBY S5A disconnects the h.t. from all stages
and connects R119A across the h.t. as a
compensating load.

(2) MANUAL (a) The h.t. passed through S5A, S5B
and S5C to all stages except the
calibration unit.

(b) S5F connects h.t. to the b.f.o. when
S7 is switched on.

(c) The a.v.c. line is disconnected from
the a.v.c. diode by S5D and connected
to the R.F./I.F. GAIN control (RV1)
by S5E.

(3) (a) (2) (a) and (2) (b) are applicable.

(b) S5D connects the a.v.c. line to the
a.v.c. diode.

(4) (a) H.T. is applied via S5A, S5B and S5F
to all stages except:-

The r.f. amplifier (V3)
The first v.f.o. (V5)
The first mixer (V7)
The second mixer (V9)
The final 37.5 Mc/s amplifier (V10)
The b.f.o.

(5) CHECK B.F.O. (a) (4)(a) applicable except that h.t. is
also applied to the b.f.o. via S7.

(b) (3)(b) applicable.


46. The "S" meter is connected between the cathode of V14 and a point of
preset (RV4) positive potential. It is calibrated to provide an
indication of signal strengh; a 1µV signal provides a typical reading of
between "S1" and "S3" and ascendinc "S" points in approximately 4 dB steps.
The variation in treshold is dependentupon the gain of the r.f. stages. It
should be remembered that only with the R.F./I.F. GAIN control at maximum is
the correct calibration maintained.